CHUieeemwcl06.pdf

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IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 16, NO. 6, JUNE 2006

W-Band Waveguide-Packaged InP

HEMT Reflection Grid Amplifier

Younkyu Chung

, Member, IEEE

, Chun-Tung Cheung, Michael P. DeLisio

, Senior Member, IEEE

, and

David B. Rutledge

, Fellow, IEEE

Abstract—

This letter presents a 79-GHz broadband reflec-

tion-type grid amplifier using spatial power combining to combine

the power of 64 unit cells. Each unit cell uses a two-stage cascade

configuration with InP HEMTs arranged as a differential pair.

A broadband orthogonal mode transducer (OMT) separates two

orthogonally polarized input and output signals over a 75 to

85 GHz range. In conjunction with the OMT, a mode converter

with quadruple-ridged apertures was designed to enhance the

field uniformity over the active grid. Measurements show 5-dB

small signal gain at 79 GHz and an 800-MHz 3-dB bandwidth.

The amplifier generates an output power of 264 mW with little

evidence of saturation.

Index Terms—

Grid amplifier, mode converter, orthogonal mode

transducer (OMT), spatial power combining.

I. I

NTRODUCTION

PATIAL power combining techniques have shown potential

for high power applications at high operating frequencies

[1]. This is essentially due to the capability to combine many

solid-state devices in free space. Large-scale power combining

is difficult using conventional power combining schemes be-

cause of inevitable loss through signal lines [2].

Recently, high-performance grid amplifiers using spatial

power combining have been extensively investigated and

reported [3]–[5]. Grid amplifiers have been realized as both

transmission and reflection type. The former has separate input

and output signal transmission planes, while those signals

coexist in the same plane in the latter [3]. DeLisio

et al.

demon-

strated a 31-GHz two-stage transmission grid amplifier module

and over 10 W output power and 12-dB gain [4]. Cheung

al.

presented 82-GHz reflection grid amplifier with 5.5-dB

gain, 400-MHz 3-dB bandwidth, and 110-mW saturated output

power [5]. While reflection type amplifiers have excellent

thermal properties, they often suffer from narrow bandwidths.

This is because the input and output are in the same trans-

mission path so that both the input and the output matching

structures must be shared.

Manuscript received November 14, 2005; revised February 27, 2006. This

work was supported by the Lee Center for Advanced Networking at California

Institute of Technology.

Y. Chung was with Information Science and Technology, California Insti-

tute of Technology (Caltech), Pasadena, CA 91125 USA and he is now with

Freescale Semiconductor Inc., Tempe, AZ 85284 USA (e-mail: ykchung@ieee.

org).

C.-T. Cheung and M. P. DeLisio are with Wavestream Corporation, San

Dimas, CA 91773 USA.

D. B. Rutledge is with the California Institute of Technology (Caltech),

Pasadena, CA 91125 USA (e-mail: rutledge@caltech.edu).

Digital Object Identifier 10.1109/LMWC.2006.875629

Fig. 1. Simplified RF schematic of bisected circuit of the differentially-driven

cascade transistor pair in the unit-cell.

A W-band reflection grid amplifier with 264-mW unsaturated

output power and 0.8-GHz 3-dB bandwidth was developed. A

broadband orthogonal mode transducer (OMT) and mode con-

verter with quadruple ridged apertures was employed to im-

prove the frequency bandwidth.

II. C

IRCUIT

ESIGN

A. Two-Stage Cascade Unit Cell

Due to the relatively small gain of a single-stage unit cell

at millimeter-wave frequencies, a two-stage cascade was em-

ployed and designed for the unit cell of the grid amplifier. Each

unit cell consists of a pair of differentially-driven transistors,

gate and drain leads, and bias circuitry. In Fig. 1 the bisected

circuit of the differential transistor pair is shown.

The drain bias voltage of the first stage is applied through the

feedback resistor

. The gate voltage of the second tran-

sistor is self-biased by means of the shunt bias resistor

The interstage matching between the two transistors is essen-

tially accomplished by elements

, and

. The

(

and

) parallel circuit is used for the input impedance-

transformation to 20

. The circuit was stabilized by

and

The design of the gate and drain leads connected to the tran-

sistor pairs was done by following the modeling procedures de-

scribed in [6]. The gate peripheries of the first and second stage

HEMTs are 80

m and 150

m, respectively. An 8

8 array

of the unit cells with a 5.5 mm

5 mm total size was used for

the final grid chip. The designed grid was fabricated by using

Northrop Grumman Space Technology’s 0.15-

m InP HEMT

process, which is used in [5].

B. Waveguide OMT and Mode Converter

The passive waveguide in this work is essentially composed

of three functional sections illustrated in Fig. 2: an OMT, a mode

converter, and an impedance matching section for the active grid

CHUNG

et al.

: W-BAND WAVEGUIDE-PACKAGED INPHEMT REFLECTION GRID AMPLIFIER

325

Fig. 2. Illustration of the waveguides in HFSS simulation including the OMT,

mode converter, matching parts. This is the symmetrical cross section along the

polarization (

-plane).

chip. Ansoft

’

s HFSS full-wave simulator was used for the de-

sign of those waveguide parts.

Since both orthogonally polarized input and output signals

coexist in the same plane, the OMT is very important. As a three-

port device, the OMT separates two orthogonally polarized sig-

nals into two standard waveguide ports, while providing a low

cross-coupling, a high port-isolation, a low transmission loss,

and low input and output return losses. The design speci

cations

of the waveguide section with the OMT and mode converter for

the input/output matching and port-isolation to be better than

10 dB and

30 dB, respectively, over the 75

–

85 GHz fre-

quency range.

Many unit cells are incorporated in the grid ampli

er so that

the size of the array is larger than that of single-mode waveg-

uides. This requires a mode converter which transforms a stan-

dard waveguide to an over-moded one while avoiding any exci-

tation of undesired modes. The third port of the OMT, as shown

in Fig. 2, was expanded to the over-moded waveguide through

the mode converter. The main role of the mode converter is to

obtain selected modes (

and

) with proper magnitude

and phase so that the

eld

atness over the aperture of the ac-

tive grid can be maximized. Furthermore the mode converter

is required to suppress undesired higher order modes such as

and

and to provide low input and output return loss.

In this work, three compact intermediate waveguide steps were

used to expand the aperture size to 0.24 in

0.24 in. Symmet-

rical quadruple ridged-type waveguide sections were used for

enhancing frequency bandwidth. The suppression goal of the

OMT and mode converter was set to

20 dB below for the

and

modes at target frequencies.

The other section of the waveguide is for impedance

matching. The grid chip was modeled as a 20-

2-D impedance

boundary on top of the InP substrate in the HFSS simulation.

The 10-mil-thick Aluminum Nitride (AlN) is used as a heat

spreader as well as a tuning element; it is placed under the

gird against the back short. In addition to the AlN substrate,

the impedance matching was accomplished by placing an

additional AlN quarter-wave length transformer and an iris as

shown in Fig. 2.

Fig. 3. Measured (

) and HFSS-simulated (

)

-parameters of the

OMT in conjunction with mode converter over 75 to 85 GHz.

III. M

EASUREMENT

ESULTS

The designed waveguides described in Section II were fabri-

cated out of brass and gold plated to reduce metallic loss. Due

to the complexity of the OMT, it was split into two blocks along

the center of

input polarization and combined together.

Each waveguide step of the mode converter was built separately

and all fabricated sections were stacked together. The measured

-parameters of the implemented OMT with the mode converter

are shown in Fig. 3. The measured return loss for the input and

output ports and the port-isolation agree well with the simu-

lation results. The transmission loss between the input/output

ports and the end of the mode converter is less than 0.5 dB over

the design frequency.

To assemble active grid chip, a 10-mil-thick AlN heat

spreader was

rst mounted on top of the brass back short. Then,

the chip was mounted on top of the AlN substrate by means of

a non-conductive epoxy with a high thermal conductivity. Gold

wires were bonded to provide dc bias. For the drain (second

stage) and gate (

rst stage) bias circuits, microstrip low-pass

lters with nulls around 79 GHz are used to prevent RF signal

leakage to dc supply ports. The bias circuitry was printed on

10-mil-thick RT/Duroid substrate with a dielectric constant of

2.2. Fig. 4(a) shows a photograph of the mounted grid ampli

chip with bias circuitry. Then, both the mounted active grid

and passive waveguide sections including the OMT, mode

converter, and matching section were combined together. The

assembled grid ampli

er with an air-cooled heat sink is shown

in Fig. 4(b).

The small-signal performance of the grid ampli

er was tested

rst. The measured

-parameters over 77 to 81 GHz are shown

in Fig. 5. The peak gain of about 5 dB at 78.8 GHz was achieved

with the

1.8 V and

0.3 V. As shown in Fig. 5,

a 3-dB-gain bandwidth of 800 MHz was obtained. Compared

to the 400-MHz bandwidth of [5], the bandwidth of the im-

plemented grid was signi

cantly expanded due to the OMT,

mode converter, and matching. The input and output return loss

below

10 dB was achieved over a frequency range from 78.5

to 79.5 GHz. Note that there was no oscillation observed in

326

IEEE MICROWAVE AND WIRELESS COMPONENTS LETTERS, VOL. 16, NO. 6, JUNE 2006

Fig. 4. (a) Photograph of mounted grid chip. (b) Photograph of assembled grid

ampli

er.

Fig. 5. Measured small signal

S

-parameters of the assembled grid ampli

er.

the measurement because the OMT suppresses the coupling be-

tween the input and output signals.

A driver module was used for the input source of the grid

ampli

er in large-signal measurement. The maximum output

power of the driver at 79 GHz is limited to 20 dBm in the mea-

surement, which was not suf

cient to saturate our ampli

er. In

Fig. 6, the measured output power and gain with respect to the

applied input power is shown. Note that the

was set to 2.2 V

and dc drain current was 9.1 A in the measurement. The unsat-

urated maximum output power of 24.2 dBm (264 mW) with a

maximum gain of 5.2 dB. Note that the ideal power density per

Fig. 6. Measured large-signal performance of the grid ampli

er with respect

to the input power.

gate periphery of the HEMT is 300 mW/mm [5]. As clearly seen

in Fig. 6, the implemented grid ampli

er does not show output

power-saturation behavior up to the maximum attainable input

power level. The output power can be increased with a higher

power driver. The gain variation may be caused by a weak non-

linear behavior of the unit cells, resulting in the gain expansion

of the gird ampli

er at the input power level higher than 12 dBm.

IV. C

ONCLUSION

A W-band re

ection grid ampli

er with an 800-MHz 3-dB-

gain bandwidth is demonstrated. A 260-mW unsaturated output

power with 5-dB small-signal gain was achieved from the im-

plemented grid ampli

er.

CKNOWLEDGMENT

The authors would like to thank Dr. J. Kuno and J. Ma, Quin-

star technology Inc., for their generous help with the large-signal

measurement, R. Tsai, Northrop Grumman, for fabrication of

the grid chips, and Dr. S. Weinreb, Jet Propulsion Laboratory,

for his helpful discussions.

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